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LT1956/LT1956-5 High Voltage, 1.5A, 500kHz Step-Down Switching Regulators
FEATURES
s s s
DESCRIPTIO
s s
s s s s s s s
Wide Input Range: 5.5V to 60V 1.5A Peak Switch Current Small 16-Pin SSOP or Thermally Enhanced TSSOP Package Saturating Switch Design: 0.2 Peak Switch Current Maintained Over Full Duty Cycle Range Constant 500kHz Switching Frequency Effective Supply Current: 2.5mA Shutdown Current: 25A 1.2V Feedback Reference (LT1956) 5V Fixed Output (LT1956-5) Easily Synchronizable Cycle-by-Cycle Current Limiting
The LT (R)1956/LT1956-5 are 500kHz monolithic buck switching regulators with an input voltage capability up to 60V. A high efficiency 1.5A, 0.2 switch is included on the die along with all the necessary oscillator, control and logic circuitry. A current mode architecture provides fast transient response and good loop stability. Special design techniques and a new high voltage process achieve high efficiency over a wide input range. Efficiency is maintained over a wide output current range by using the output to bias the circuitry and by utilizing a supply boost capacitor to saturate the power switch. Patented circuitry maintains peak switch current over the full duty cycle range*. A shutdown pin reduces supply current to 25A and the device can be externally synchronized from 580kHz to 700kHz with a logic level input. The LT1956/LT1956-5 are available in fused-lead 16-pin SSOP and thermally enhanced TSSOP packages.
, LTC and LT are registered trademarks of Linear Technology Corporation. *U.S. PATENT NO. 6,498,466
APPLICATIO S
s s s s s
High Voltage, Industrial and Automotive Portable Computers Battery-Powered Systems Battery Chargers Distributed Power Systems
TYPICAL APPLICATIO
5V Buck Converter
MMSD914TI 6 VIN 12V (TRANSIENTS TO 60V) BOOST 4 2.2F 100V CERAMIC 15 14 VIN LT1956-5 SHDN SYNC GND BIAS FB VC 10 12 SW 2 0.1F 10H 10MQ060N VOUT 5V 1A 22F 6.3V CERAMIC
EFFICIENCY (%)
1, 8, 9, 16 11
220pF 4.7k 4700pF
UNITED CHEMI-CON THCS50EZA225ZT
1956 TA01
U
Efficiency vs Load Current
100 VIN = 12V L = 18H 90 VOUT = 5V VOUT = 3.3V 80 70 60 50 0 0.25 0.75 1.00 0.50 LOAD CURRENT (A) 1.25
1956 TA02
U
U
1956f
1
LT1956/LT1956-5
ABSOLUTE
AXI U
RATI GS
Input Voltage (VIN) ................................................. 60V BOOST Pin Above SW ............................................ 35V BOOST Pin Voltage ................................................. 68V SYNC, SENSE Voltage (LT1956-5) ........................... 7V SHDN Voltage ........................................................... 6V BIAS Pin Voltage .................................................... 30V FB Pin Voltage/Current (LT1956) ................... 3.5V/2mA
PACKAGE/ORDER I FOR ATIO
TOP VIEW GND SW NC VIN NC BOOST NC GND 1 2 3 4 5 6 7 8 16 GND 15 SHDN 14 SYNC 13 NC 12 FB/SENSE 11 VC 10 BIAS 9 GND
ORDER PART NUMBER LT1956EFE LT1956IFE LT1956EFE-5 LT1956IFE-5 FE PART MARKING 1956EFE 1956IFE 1956EFE-5 1956IFE-5
FE PACKAGE 16-LEAD PLASTIC TSSOP
TJMAX = 125C, JA = 45C/ W, JC (PAD) = 10C/ W EXPOSED BACKSIDE MUST BE SOLDERED TO GROUND PLANE
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25C. VIN = 15V, VC = 1.5V, SHDN = 1V, Boost o/c, SW o/c, unless otherwise noted.
PARAMETER Reference Voltage (LT1956) SENSE Voltage (LT1956-5) SENSE Pin Resistance (LT1956-5) FB Input Bias Current (LT1956) Error Amp Voltage Gain Error Amp gm VC to Switch gm EA Source Current EA Sink Current VC Switching Threshold VC High Clamp FB = 1V or VSENSE = 4.1V FB = 1.4V or VSENSE = 5.7V Duty Cycle = 0 SHDN = 1V
q q q
CONDITIONS 5.5V VIN 60V VOL + 0.2 VC VOH - 0.2 5.5V VIN 60V VOL + 0.2 VC VOH - 0.2
q q
(Notes 2, 9) dl (VC) = 10A (Note 9)
q
2
U
U
W
WW U
W
(Note 1)
Operating Junction Temperature Range LT1956EFE/LT1956EFE-5/LT1956EGN/LT1956EGN-5 (Notes 8, 10) ..................................... - 40C to 125C LT1956IFE/LT1956IFE-5/LT1956IGN/LT1956IGN-5 (Notes 8, 10) ..................................... - 40C to 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
TOP VIEW GND SW NC VIN NC BOOST NC GND 1 2 3 4 5 6 7 8 16 GND 15 SHDN 14 SYNC 13 NC 12 FB/SENSE 11 VC 10 BIAS 9 GND
ORDER PART NUMBER LT1956EGN LT1956IGN LT1956EGN-5 LT1956IGN-5 GN PART MARKING 1956 1956I 19565 1956I5
GN PACKAGE 16-LEAD PLASTIC SSOP
TJMAX = 125C, JA = 85C/ W, JC (PIN 8) = 25C/ W FOUR CORNER PINS SOLDERED TO GROUND PLANE
MIN 1.204 1.195 4.94 4.90 9.5 200 1500 1000 125 100
TYP 1.219 5 13.8 0.5 400 2000 1.7 225 225 0.9 2.1
MAX 1.234 1.243 5.06 5.10 19 1.5 3000 3200 400 450
UNITS V V V V k A V/V Mho Mho A/V A A V V
1956f
LT1956/LT1956-5
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25C. VIN = 15V, VC = 1.5V, SHDN = 1V, Boost o/c, SW o/c, unless otherwise noted.
PARAMETER Switch Current Limit Switch On Resistance Maximum Switch Duty Cycle Switch Frequency fSW Line Regulation fSW Shifting Threshold Minimum Input Voltage Minimum Boost Voltage Boost Current (Note 5) Input Supply Current (IVIN) Output Supply Current (IBIAS) Shutdown Supply Current Lockout Threshold Shutdown Thresholds Minimum SYNC Amplitude SYNC Frequency Range SYNC Input Resistance Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Gain is measured with a VC swing equal to 200mV above the low clamp level to 200mV below the upper clamp level. Note 3: Minimum input voltage is not measured directly, but is guaranteed by other tests. It is defined as the voltage where internal bias lines are still regulated so that the reference voltage and oscillator remain constant. Actual minimum input voltage to maintain a regulated output will depend upon output voltage and load current. See Applications Information. Note 4: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch. Note 5: Boost current is the current flowing into the BOOST pin with the pin held 5V above input voltage. It flows only during switch on time. Note 6: Input supply current is the quiescent current drawn by the input pin when the BIAS pin is held at 5V with switching disabled. Bias supply current is the current drawn by the BIAS pin when the BIAS pin is held at 5V. Total input referred supply current is calculated by summing input supply current (IVIN) with a fraction of supply current (IBIAS): ITOTAL = IVIN + (IBIAS)(VOUT/VIN) with VIN = 15V, VOUT = 5V, IVIN = 1.4mA, IBIAS = 2.9mA, ITOTAL = 2.4mA. CONDITIONS VC Open, Boost = VIN + 5V, FB = 1V or VSENSE = 4.1V q ISW = 1.5A, Boost = VIN + 5V (Note 7)
q
MIN 1.5
TYP 2 0.2
MAX 3 0.3 0.4
UNITS A % %
FB = 1V or VSENSE = 4.1V
q
82 75 460 430
90 90 500 0.05 0.8 540 570 0.15
VC Set to Give DC = 50%
q
kHz kHz %/V V
5.5V VIN 60V Df = 10kHz (Note 3) (Note 4) ISW 1.5A Boost = VIN + 5V, ISW = 0.5A Boost = VIN + 5V, ISW = 1.5A (Note 6) VBIAS = 5V (Note 6) VBIAS = 5V SHDN = 0V, VIN 60V, SW = 0V, VC Open
q
q q q q
4.6 2 12 42 1.4 2.9 25
5.5 3 25 70 2.2 4.2 75 200 2.53 0.6 0.6 2.2 700
V V mA mA mA mA A A V V V V kHz k
q
VC Open VC Open, Shutting Down VC Open, Starting Up
q q q q
2.30 0.15 0.25 580
2.42 0.37 0.45 1.5 20
Note 7: Switch on resistance is calculated by dividing VIN to SW voltage by the forced current (1.5A). See Typical Performance Characteristics for the graph of switch voltage at other currents. Note 8: The LT1956EFE/LT1956EFE-5/LT1956EGN/LT1956EGN-5 are guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT1956IFE/LT1956IFE-5/ LT1956IGN/LT1956IGN-5 are guaranteed over the full - 40C to 125C operating junction temperature range. Note 9: Transconductance and voltage gain refer to the internal amplifier exclusive of the voltage divider. To calculate gain and transconductance, refer to the SENSE pin on fixed voltage parts. Divide values shown by the ratio VOUT/1.219. Note 10: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
1956f
3
LT1956/LT1956-5 TYPICAL PERFOR A CE CHARACTERISTICS
Switch Peak Current Limit
2.5
1.234 1.229
SWITCH PEAK CURRENT (A)
TYPICAL 2.0
FEEDBACK VOLTAGE (V)
VOLTAGE 1.219 CURRENT 1.214 0.5 1.209 1.0
CURRENT (A)
1.5
GUARANTEED MINIMUM
1.0 0 20 40 60 DUTY CYCLE (%) 80 100
1956 G01
Lockout and Shutdown Thresholds
2.4
INPUT SUPPLY CURRENT (A)
40
INPUT SUPPLY CURRENT (A)
2.0 SHDN PIN VOLTAGE (V) 1.6 1.2 0.8
LOCKOUT
START-UP 0.4 SHUTDOWN 0 -50
-25
0
25
50
75
JUNCTION TEMPERATURE (C)
1956 G04
Error Amplifier Transconductance
2500
TRANSCONDUCTANCE (mho)
2000
GAIN (Mho)
2500 GAIN
150
PHASE (DEG)
SWITICHING FREQUENCY (kHz) OR FB CURRENT (A)
1500
1000
500
0 -50
-25
0
25
50
75
JUNCTION TEMPERATURE
1956 G07
4
UW
100 100
FB Pin Voltage and Current
2.0 250 200 1.5 1.224 150 100 12 6 0 125
SHDN Pin Bias Current
CURRENT REQUIRED TO FORCE SHUTDOWN (FLOWS OUT OF PIN). AFTER SHUTDOWN, CURRENT DROPS TO A FEW A
CURRENT (A)
AT 2.38V STANDBY THRESHOLD (CURRENT FLOWS OUT OF PIN)
1.204 50 100 25 75 -50 -25 0 JUNCTION TEMPERATURE (C)
0 50 100 -50 -25 25 75 0 JUNCTION TEMPERATURE (C)
125
1956 G02
1956 G03
Shutdown Supply Current
300
VSHDN = 0V 35 30 25 20 15 10 5 0 0 10 20 30 40 INPUT VOLTAGE (V) 50 60
1956 G05
Shutdown Supply Current
250 VIN = 60V 200 VIN = 15V 150 100 50 0 0 0.1 0.2 0.3 0.4 SHUTDOWN VOLTAGE (V) 0.5
1956 G06
125
Error Amplifier Transconductance
3000 PHASE 200
Frequency Foldback
625 SWITCHING FREQUENCY
500
2000
100
VC
375
1500
VFB 2 * 10-3
(
)
ROUT 200k
COUT 12pF
50
250
1000
ERROR AMPLIFIER EQUIVALENT CIRCUIT
0
125 FB PIN CURRENT 0 0 0.2 0.4 0.6 VFB (V) 0.8 1.0 1.2
1956 G09
RLOAD = 50
125
500 100
1k
10k 100k FREQUENCY (Hz)
1M
-50 10M
1956 G08
1956f
LT1956/LT1956-5 TYPICAL PERFOR A CE CHARACTERISTICS
Switching Frequency
575 550 7.5
BOOST PIN CURRENT (mA)
INPUT VOLTAGE (V)
FREQUENCY (kHz)
525 500 475 450 425 -50
-25
0
25
50
75
JUNCTION TEMPERATURE (C)
1956 G10
VC Pin Shutdown Threshold
2.1 1.9
THRESHOLD VOLTAGE (V)
450 400
TJ = 125C
SWITCH MINIMUM ON TIME (ns)
1.7 1.5 1.3 1.1 0.9 0.7 50 100 -50 -25 25 75 0 JUNCTION TEMPERATURE (C)
SWITCH VOLTAGE (mV)
UW
100
1956 G13
Minimum Input Voltage with 5V Output
VOUT = 5V L = 18H 45 40 35 30 25 20 15 10 5 125 5.0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 LOAD CURRENT (A) 1 0
BOOST Pin Current
7.0 MINIMUM INPUT VOLTAGE TO START
6.5
6.0 MINIMUM INPUT VOLTAGE TO RUN 5.5
0
0.5 1 SWITCH CURRENT (A)
1.5
1956 G12
1956 G11
Switch Voltage Drop
600 500 400 300 200 100
Switch Minimum ON Time vs Temperature
350 300 250 200 150 100 50 TJ = -40C TJ = 25C
125
0
0
0.5 1 SWITCH CURRENT (A)
1.5
1766 G14
0 -50
-25
0
25
50
75
100
125
JUNCTION TEMPERATURE (C)
1956 G15
1956f
5
LT1956/LT1956-5
PI FU CTIO S
GND (Pins 1, 8, 9, 16): The GND pin connections act as the reference for the regulated output, so load regulation will suffer if the "ground" end of the load is not at the same voltage as the GND pins of the IC. This condition will occur when load current or other currents flow through metal paths between the GND pins and the load ground. Keep the paths between the GND pins and the load ground short and use a ground plane when possible. For the FE package, the exposed pad should be soldered to the copper GND plane underneath the device. (See Applications Information--Layout Considerations.) SW (Pin 2): The switch pin is the emitter of the on-chip power NPN switch. This pin is driven up to the input pin voltage during switch on time. Inductor current drives the switch pin negative during switch off time. Negative voltage is clamped with the external catch diode. Maximum negative switch voltage allowed is - 0.8V. NC (Pins 3, 5, 7, 13): No Connection. VIN (Pin 4): This is the collector of the on-chip power NPN switch. VIN powers the internal control circuitry when a voltage on the BIAS pin is not present. High dI/dt edges occur on this pin during switch turn on and off. Keep the path short from the VIN pin through the input bypass capacitor, through the catch diode back to SW. All trace inductance on this path will create a voltage spike at switch off, adding to the VCE voltage across the internal NPN. BOOST (Pin 6): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. Without this added voltage, the typical switch voltage loss would be about 1.5V. The additional BOOST voltage allows the switch to saturate and voltage loss approximates that of a 0.2 FET structure, but with much smaller die area. BIAS (Pin 10): The BIAS pin is used to improve efficiency when operating at higher input voltages and light load current. Connecting this pin to the regulated output voltage forces most of the internal circuitry to draw its operating current from the output voltage rather than the input supply. This architecture increases efficiency especially when the input voltage is much higher than the output. Minimum output voltage setting for this mode of operation is 3V. VC (Pin 11) The VC pin is the output of the error amplifier and the input of the peak switch current comparator. It is normally used for frequency compensation, but can also serve as a current clamp or control loop override. VC sits at about 1V for light loads and 2V at maximum load. It can be driven to ground to shut off the regulator, but if driven high, current must be limited to 4mA. FB/SENSE (Pin 12): The feedback pin is used to set the output voltage using an external voltage divider that generates 1.22V at the pin for the desired output voltage. The 5V fixed output voltage parts have the divider included on the chip and the FB pin is used as a SENSE pin, connected directly to the 5V output. Three additional functions are performed by the FB pin. When the pin voltage drops below 0.6V, switch current limit is reduced and the external SYNC function is disabled. Below 0.8V, switching frequency is also reduced. See Feedback Pin Functions in Applications Information for details. SYNC (Pin 14): The SYNC pin is used to synchronize the internal oscillator to an external signal. It is directly logic compatible and can be driven with any signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency up to 700kHz. See Synchronizing in Applications Information for details. If unused, this pin should be tied to ground. SHDN (Pin 15): The SHDN pin is used to turn off the regulator and to reduce input current to a few microamperes. This pin has two thresholds: one at 2.38V to disable switching and a second at 0.4V to force complete micropower shutdown. The 2.38V threshold functions as an accurate undervoltage lockout (UVLO); sometimes used to prevent the regulator from delivering power until the input voltage has reached a predetermined level. If the SHDN pin functions are not required, the pin can either be left open (to allow an internal bias current to lift the pin to a default high state) or be forced high to a level not to exceed 6V.
6
U
U
U
1956f
LT1956/LT1956-5
BLOCK DIAGRA
The LT1956 is a constant frequency, current mode buck converter. This means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. In addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on. When switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. Output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. This technique means that the error amplifier commands current to be delivered to the output rather than voltage. A voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180 shift will occur. The current fed system will have 90 phase shift at a much lower frequency, but will not have the additional 90 shift until well beyond the LC resonant frequency. This makes
VIN 4
BIAS 10
SLOPE COMP SYNC 14 ANTISLOPE COMP SHUTDOWN COMPARATOR 500kHz OSCILLATOR
0.4V 5.5A SHDN 15
+ -
LOCKOUT COMPARATOR x1 Q2 FOLDBACK CURRENT LIMIT CLAMP FREQUENCY FOLDBACK
VC(MAX) CLAMP
Q3
2.38V
11 VC
Figure 1. LT1956 Block Diagram
1956f
-
+
-
+
2.9V BIAS REGULATOR
W
it much easier to frequency compensate the feedback loop and also gives much quicker transient response. Most of the circuitry of the LT1956 operates from an internal 2.9V bias line. The bias regulator normally draws power from the regulator input pin, but if the BIAS pin is connected to an external voltage higher than 3V, bias power will be drawn from the external source (typically the regulated output voltage). This will improve efficiency if the BIAS pin voltage is lower than regulator input voltage. High switch efficiency is attained by using the BOOST pin to provide a voltage to the switch driver which is higher than the input voltage, allowing switch to be saturated. This boosted voltage is generated with an external capacitor and diode. Two comparators are connected to the shutdown pin. One has a 2.38V threshold for undervoltage lockout and the second has a 0.4V threshold for complete shutdown.
RLIMIT INTERNAL VCC RSENSE CURRENT COMPARATOR BOOST 6 S R RS FLIP-FLOP DRIVER CIRCUITRY Q1 POWER SWITCH
-
+
2
SW
ERROR AMPLIFIER gm = 2000Mho
12 FB
1.22V GND 1, 8, 9, 16
1956 F01
7
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
FEEDBACK PIN FUNCTIONS
The feedback (FB) pin on the LT1956 is used to set output voltage and provide several overload protection features. The first part of this section deals with selecting resistors to set output voltage and the remaining part talks about foldback frequency and current limiting created by the FB pin. Please read both parts before committing to a final design. The 5V fixed output voltage part (LT1956-5) has internal divider resistors and the FB pin is renamed SENSE, connected directly to the output. The suggested value for the output divider resistor (see Figure 2) from FB to ground (R2) is 5k or less, and a formula for R1 is shown below. The output voltage error caused by ignoring the input bias current on the FB pin is less than 0.25% with R2 = 5k. A table of standard 1% values is shown in Table 1 for common output voltages. Please read the following section if divider resistors are increased above the suggested values.
R1 =
Table 1
OUTPUT VOLTAGE (V) 3 3.3 5 6 8 10 12 15 R2 (k) 4.99 4.99 4.99 4.75 4.47 4.32 4.12 4.12 R1 (NEAREST 1%) (k) 7.32 8.45 15.4 18.7 24.9 30.9 36.5 46.4 % ERROR AT OUTPUT DUE TO DISCRETE 1% RESISTOR STEPS + 0.32 - 0.43 - 0.30 + 0.38 + 0.20 - 0.54 + 0.24 - 0.27
R2( VOUT - 1.22) 1.22
More Than Just Voltage Feedback The feedback pin is used for more than just output voltage sensing. It also reduces switching frequency and current limit when output voltage is very low (see the Frequency Foldback graph in Typical Performance Characteristics). This is done to control power dissipation in both the IC and in the external diode and inductor during short-circuit conditions. A shorted output requires the switching regulator to operate at very low duty cycles, and the average
8
U
current through the diode and inductor is equal to the short-circuit current limit of the switch (typically 2A for the LT1956, folding back to less than 1A). Minimum switch on time limitations would prevent the switcher from attaining a sufficiently low duty cycle if switching frequency were maintained at 500kHz, so frequency is reduced by about 5:1 when the feedback pin voltage drops below 0.8V (see Frequency Foldback graph). This does not affect operation with normal load conditions; one simply sees a shift in switching frequency during start-up as the output voltage rises. In addition to lower switching frequency, the LT1956 also operates at lower switch current limit when the feedback pin voltage drops below 0.6V. Q2 in Figure 2 performs this function by clamping the VC pin to a voltage less than its normal 2.1V upper clamp level. This foldback current limit greatly reduces power dissipation in the IC, diode and inductor during short-circuit conditions. External synchronization is also disabled to prevent interference with foldback operation. Again, it is nearly transparent to the user under normal load conditions. The only loads that may be affected are current source loads which maintain full load current with output voltage less than 50% of final value. In these rare situations the feedback pin can be clamped above 0.6V with an external diode to defeat foldback current limit. Caution: clamping the feedback pin means that frequency shifting will also be defeated, so a combination of high input voltage and dead shorted output may cause the LT1956 to lose control of current limit. The internal circuitry which forces reduced switching frequency also causes current to flow out of the feedback pin when output voltage is low. The equivalent circuitry is shown in Figure 2. Q1 is completely off during normal operation. If the FB pin falls below 0.8V, Q1 begins to conduct current and reduces frequency at the rate of approximately 3.5kHz/A. To ensure adequate frequency foldback (under worst-case short-circuit conditions), the external divider Thevinin resistance must be low enough to pull 115A out of the FB pin with 0.44V on the pin (RDIV 3.8k). The net result is that reductions in frequency and current limit are affected by output voltage divider impedance. Although divider impedance is not critical, caution should be used if resistors are increased beyond the suggested values and short-circuit conditions will occur
1956f
W
UU
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
LT1956 ERROR AMPLIFIER
TO FREQUENCY SHIFTING 1.4V Q1
+ -
Q2 TO SYNC CIRCUIT
VC
GND
Figure 2. Frequency and Current Limit Foldback
with high input voltage. High frequency pickup will increase and the protection accorded by frequency and current foldback will decrease.
CHOOSING THE INDUCTOR For most applications, the output inductor will fall into the range of 5H to 30H. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the LT1956 switch, which has a 1.5A limit. Higher values also reduce output ripple voltage. When choosing an inductor you will need to consider output ripple voltage, maximum load current, peak inductor current and fault current in the inductor. In addition, other factors such as core and copper losses, allowable component height, EMI, saturation and cost should also be considered. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. Output Ripple Voltage Figure 3 shows a comparison of output ripple voltage for the LT1956 using either a tantalum or ceramic output capacitor. It can be seen from Figure 3 that output ripple voltage can be significantly reduced by using the ceramic output capacitor; the significant decrease in output ripple voltage is due to the very low ESR of ceramic capacitors.
U
VSW L1 OUTPUT 5V 1.2V R3 1k R4 2k BUFFER R2 FB R1
W
UU
+
C1
1956 F02
10mV/DIV
VOUT USING 22F CERAMIC OUTPUT CAPACITOR
10mV/DIV
VOUT USING 100F, 0.08 TANTALUM OUTPUT CAPACITOR VIN = 12V VOUT = 5V L = 15H 1s/DIV
1956 F03
Figure 3. LT1956 Output Ripple Voltage Waveforms. Ceramic vs Tantalum Output Capacitors
Output ripple voltage is determined by ripple current (ILP-P) through the inductor and the high frequency impedance of the output capacitor. At high frequencies, the impedance of the tantalum capacitor is dominated by its effective series resistance (ESR). Tantalum Output Capacitor The typical method for reducing output ripple voltage when using a tantalum output capacitor is to increase the inductor value (to reduce the ripple current in the inductor). The following equations will help in choosing the required inductor value to achieve a desirable output ripple voltage level. If output ripple voltage is of less importance, the subsequent suggestions in Peak Inductor and Fault Current and EMI will additionally help in the selection of the inductor value.
1956f
9
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
Peak-to-peak output ripple voltage is the sum of a triwave (created by peak-to-peak ripple current (ILP-P) times ESR) and a square wave (created by parasitic inductance (ESL) and ripple current slew rate). Capacitive reactance is assumed to be small compared to ESR or ESL. VRIPPLE = (ILP-P )(ESR) + (ESL) where: ESR = equivalent series resistance of the output capacitor ESL = equivalent series inductance of the output capacitor dI/dt = slew rate of inductor ripple current = VIN/L Peak-to-peak ripple current (ILP-P) through the inductor and into the output capacitor is typically chosen to be between 20% and 40% of the maximum load current. It is approximated by: ILP-P = dI dt
(VOUT )(VIN - VOUT ) (VIN )( f)(L)
Example: with VIN = 12V, VOUT = 5V, L = 15H, ESR = 0.080 and ESL = 10nH, output ripple voltage can be approximated as follows:
ILP-P =
(12)(15 * 10-6 )(500 * 10-6 )
(5)(12 - 5)
= 0.389 A
12 dI = 106 * 0.8 = -6 dt 15 * 10
VRIPPLE = (0.389)(0.08) + 10 * 10 - 9 10 6 (0.8 ) = 0.031 + 0.008 = 39mVP-P
(
)( )
To reduce output ripple voltage further requires an increase in the inductor value with the trade-off being a physically larger inductor with the possibility of increased component height and cost. Ceramic Output Capacitor An alternative way to further reduce output ripple voltage is to reduce the ESR of the output capacitor by using a
10
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ceramic capacitor. Although this reduction of ESR removes a useful zero in the overall loop response, this zero can be replaced by inserting a resistor (RC) in series with the VC pin and the compensation capacitor CC. (See Ceramic Capacitors in Applications Information.) Peak Inductor Current and Fault Current To ensure that the inductor will not saturate, the peak inductor current should be calculated knowing the maximum load current. An appropriate inductor should then be chosen. In addition, a decision should be made whether or not the inductor must withstand continuous fault conditions. If maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 2A overload condition. Dead shorts will actually be more gentle on the inductor because the LT1956 has frequency and current limit foldback. Peak inductor and switch current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don't omit this step. Powdered
Table 2
VENDOR/ PART NO. Coiltronics UP1B-100 UP1B-220 UP2B-220 UP2B-330 UP1B-150 Coilcraft D01813P-153HC D01813P-103HC D53316P-223 D53316P-333 LP025060B-682 Sumida CDRH4D28-4R7 CDRH5D28-100 CDRH6D28-150 CDRH6D28-180 CDRH6D28-220 CDRH6D38-220 4.7 10 15 18 22 22 1.32 1.30 1.40 1.32 1.20 1.30 0.072 0.065 0.084 0.095 0.128 0.096 3.0 3.0 3.0 3.0 3.0 4.0
1956f
W
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VALUE (H) 10 22 22 33 15 15 10 22 33 6.8
IDC(MAX) (Amps) 1.9 1.2 2.0 1.7 1.5 1.5 1.9 1.6 1.4 1.3
DCR (Ohms) 0.111 0.254 0.062 0.092 0.175 0.170 0.111 0.207 0.334 0.165
HEIGHT (mm) 5.0 5.0 6.0 6.0 5.0 5.0 5.0 5.1 5.1 1.65
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iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall somewhere in between. The following formula assumes continuous mode of operation, but errs only slightly on the high side for discontinuous mode, so it can be used for all conditions.
IPEAK = IOUT +
EMI
VOUT (VIN - VOUT ) ILP-P = IOUT + 2 2 * VIN * f * L
Decide if the design can tolerate an "open" core geometry like a rod or barrel, which have high magnetic field radiation, or whether it needs a closed core like a toroid to prevent EMI problems. This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. Additional Considerations After making an initial choice, consider additional factors such as core losses and second sourcing, etc. Use the experts in Linear Technology's Applications department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. MAXIMUM OUTPUT LOAD CURRENT Maximum load current for a buck converter is limited by the maximum switch current rating (IP). The current rating for the LT1956 is 1.5A. Unlike most current mode converters, the LT1956 maximum switch current limit does not fall off at high duty cycles. Most current mode converters suffer a drop off of peak switch current for duty cycles above 50%. This is due to the effects of slope compensation required to prevent subharmonic oscillations in current mode converters. (For detailed analysis, see Application Note 19.) The LT1956 is able to maintain peak switch current limit over the full duty cycle range by using patented circuitry to cancel the effects of slope compensation on peak switch
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current without affecting the frequency compensation it provides. Maximum load current would be equal to maximum switch current for an infinitely large inductor, but with finite inductor size, maximum load current is reduced by one half of peak-to-peak inductor current (ILP-P). The following formula assumes continuous mode operation, implying that the term on the right is less than one half of IP. IOUT (MAX) Continuous Mode = IP -
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(VOUT + VF )(VIN - VOUT - VF ) ILP-P = IP - 2 (2)(VIN )( f)(L)
For VOUT = 5V, VIN(MAX) = 8V, VF(DI) = 0.63V, f = 500kHz and L = 10H: IOUT (MAX) = 1.5 -
(5 + 0.63)(8 - 5 - 0.63) (2)(8)(500 * 103 )(10 * 10-6 )
= 1.5 - 0.17 = 1.33A Note that there is less load current available at the higher input voltage because inductor ripple current increases. At VIN = 15V and using the same set of conditions: IOUT (MAX) = 1.5 -
(5 + 0.63)(15 - 5 - 0.63) (2)(15)(500 * 103 )(10 * 10-6 )
= 1.5 - 0.35 = 1.15A To calculate peak switch current with a given set of conditions, use: ISW(PEAK) = IOUT + = IOUT ILP-P 2 VOUT + VF )(VIN - VOUT - VF ) ( + (2)(VIN )( f)(L)
Reduced Inductor Value and Discontinuous Mode If the smallest inductor value is of the most importance to a converter design, in order to reduce inductor size/cost, discontinuous mode may yield the smallest inductor
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solution. The maximum output load current in discontinuous mode, however, must be calculated and is defined later in this section. Discontinuous mode is entered when the output load current is less than one-half of the inductor ripple current (ILP-P). In this mode, inductor current falls to zero before the next switch turn-on (see Figure 8). Buck converters will be in discontinuous mode for output load current given by:
IOUT Discontinous Mode (V + V )( V - V -V ) < OUT F IN OUT F (2)( VIN )( f)(L)
The inductor value in a buck converter is usually chosen large enough to keep inductor ripple current (ILP-P) low; this is done to minimize output ripple voltage and maximize output load current. In the case of large inductor values, as seen in the equation above, discontinuous mode will be associated with "light loads." When choosing small inductor values, however, discontinuous mode will occur at much higher output load currents. The limit to the smallest inductor value that can be chosen is set by the LT1956 peak switch current (IP) and the maximum output load current required given by: IOUT(MAX) Discontinuous Mode IP2 IP2 ( f)(L)(VIN ) = = 2(ILP-P ) 2( VOUT + VF )(VIN - VOUT - VF ) Example: For VIN = 15V, VOUT = 5V, VF = 0.63V, f = 500kHz and L = 4H
IOUT (MAX) Discontinuous Mode = 1.52 (500 * 103 )(4 * 10-6 )(15) 2(5 + 0.63)(15 - 5 - 0.63)
IOUT(MAX) Discontinuous Mode = 0.639A What has been shown here is that if high inductor ripple current and discontinuous mode operation can be tolerated, small inductor values can be used. If a higher output
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load current is required, the inductor value must be increased. If IOUT(MAX) no longer meets the discontinuous mode criteria, use the IOUT(MAX) equation for continuous mode; the LT1956 is designed to operate well in both modes of operation, allowing a large range of inductor values to be used. SHORT-CIRCUIT CONSIDERATIONS For a ground short-circuit fault on the regulated output, the maximum input voltage for the LT1956 is typically limited to 25V. If a greater input voltage is required, increasing the resistance in series with the inductor may suffice (see short-circuit calculations at the end of this section). Alternatively, the 1.5A LT1766 can be used since it is identical to the LT1956 but runs at a lower frequency of 200kHz, allowing higher sustained input voltage capability during output short circuit. The LT1956 is a current mode controller. It uses the VC node voltage as an input to a current comparator which turns off the output switch on a cycle-by-cycle basis as peak switch current is reached. The internal clamp on the VC node, nominally 2V, then acts as an output switch peak current limit. This action becomes the switch current limit specification. The maximum available output power is then determined by the switch current limit. A potential controllability problem could occur under short-circuit conditions. If the power supply output is short circuited, the feedback amplifier responds to the low output voltage by raising the control voltage, VC, to its peak current limit value. Ideally, the output switch would be turned on, and then turned off as its current exceeded the value indicated by VC. However, there is finite response time involved in both the current comparator and turnoff of the output switch. These result in a minimum on time tON(MIN). When combined with the large ratio of VIN to (VF + I * R), the diode forward voltage plus inductor I * R voltage drop, the potential exists for a loss of control. Expressed mathematically the requirement to maintain control is:
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f * tON
VF + I * R VIN
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where: f = switching frequency tON = switch minimum on time VF = diode forward voltage VIN = input voltage I * R = inductor I * R voltage drop
If this condition is not observed, the current will not be limited at IPK, but will cycle-by-cycle ratchet up to some higher value. Using the nominal LT1956 clock frequency of 500KHz, a VIN of 12V and a (VF + I * R) of say 0.7V, the maximum tON to maintain control would be approximately 116ns, an unacceptably short time. The solution to this dilemma is to slow down the oscillator when the FB pin voltage is abnormally low thereby indicating some sort of short-circuit condition. Oscillator frequency is unaffected until FB voltage drops to about 2/3 of its normal value. Below this point the oscillator frequency decreases roughly linearly down to a limit of about 100kHz. This lower oscillator frequency during short-circuit conditions can then maintain control with the effective minimum on time. Even with frequency foldback, however, the LT1956 will not survive a permanent output short at the absolute maximum voltage rating of VIN = 60V; this is defined solely by internal semiconductor junction breakdown effects. For the maximum input voltage allowed during an output short to ground, the previous equation defining minimum on-time can be used. Assuming VF (D1 catch diode) = 0.63V at 1A (short-circuit current is folded back to typical switch current limit * 0.5), I (inductor) * DCR = 1A * 0.128 = 0.128V (L = CDRH6D28-22), typical f = 100kHz (folded back) and typical minimum on-time = 300ns, the maximum allowable input voltage during an output short to ground is typically: VIN = (0.63V + 0.128V)/(100kHz * 300ns) VIN(MAX) = 25V Increasing the DCR of the inductor will increase the maximum VIN allowed during an output short to ground but will also drop overall efficiency during normal operation. Every time the converter wakes up from shutdown or undervoltage lockout to begin switching, the output
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capacitor may potentially be starting from 0V. This requires that the part obey the overall duty cycle demanded by the loop, related to VIN and VOUT, as the output voltage rises to its target value. It is recommended that for [VIN/ (VOUT + VF)] ratios > 4, a soft-start circuit should be used to control the output capacitor charge rate during start-up or during recovery from an output short circuit, thereby adding additional control over peak inductor current. See Buck Converter with Adjustable Soft-Start later in this data sheet. OUTPUT CAPACITOR The LT1956 will operate with either ceramic or tantalum output capacitors. The output capacitor is normally chosen by its effective series resistance (ESR), because this is what determines output ripple voltage. The ESR range for typical LT1956 applications using a tantalum output capacitor is 0.05 to 0.2. A typical output capacitor is an AVX type TPS, 100F at 10V, with a guaranteed ESR less than 0.1. This is a "D" size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR, so they give the lowest ESR for a given volume. The value in microfarads is not particularly critical, and values from 22F to greater than 500F work well, but you cannot cheat mother nature on ESR. If you find a tiny 22F solid tantalum capacitor, it will have high ESR, and output ripple voltage will be terrible. Table 3 shows some typical solid tantalum surface mount capacitors.
Table 3. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E CASE SIZE ESR (MAX, ) RIPPLE CURRENT (A)
AVX TPS, Sprague 593D D CASE SIZE AVX TPS, Sprague 593D C CASE SIZE AVX TPS 0.2 (typ) 0.5 (typ) 0.1 to 0.3 0.7 to 1.1 0.1 to 0.3 0.7 to 1.1
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Unlike the input capacitor, RMS ripple current in the output capacitor is normally low enough that ripple current rating is not an issue. The current waveform is triangular with a typical value of 125mARMS. The formula to calculate this is:
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Output capacitor ripple current (RMS): IRIPPLE(RMS) = 0.29(VOUT )(VIN - VOUT ) (L)( f)(VIN )
Ceramic Capacitors Ceramic capacitors are generally chosen for their good high frequency operation, small size and very low ESR (effective series resistance). Their low ESR reduces output ripple voltage but also removes a useful zero in the loop frequency response, common to tantalum capacitors. To compensate for this, a resistor RC can be placed in series with the VC compensation capacitor CC. Care must be taken however, since this resistor sets the high frequency gain of the error amplifier, including the gain at the switching frequency. If the gain of the error amplifier is high enough at the switching frequency, output ripple voltage (although smaller for a ceramic output capacitor) may still affect the proper operation of the regulator. A filter capacitor CF in parallel with the RC/CC network is suggested to control possible ripple at the VC pin. The LT1956 can be stabilized for VOUT = 5V at 1A using a 22F ceramic output capacitor and VC component values of CC = 4700pF, RC = 4.7k and CF = 220pF. INPUT CAPACITOR Step-down regulators draw current from the input supply in pulses. The rise and fall times of these pulses are very fast. The input capacitor is required to reduce the voltage ripple this causes at the input of LT1956 and force the switching current into a tight local loop, thereby minimizing EMI. The RMS ripple current can be calculated from: IRIPPLE(RMS)CIN = IOUT VOUT (VIN - VOUT ) VIN2
Ceramic capacitors are ideal for input bypassing. At 500kHz switching frequency, the energy storage requirement of the input capacitor suggests that values in the range of 2.2F to 10F are suitable for most applications. If operation is required close to the minimum input required by the output of the LT1956, a larger value may be required. This
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is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation. Depending on how the LT1956 circuit is powered up you may need to check for input voltage transients. The input voltage transients may be caused by input voltage steps or by connecting the LT1956 converter to an already powered up source such as a wall adapter. The sudden application of input voltage will cause a large surge of current in the input leads that will store energy in the parasitic inductance of the leads. This energy will cause the input voltage to swing above the DC level of input power source and it may exceed the maximum voltage rating of input capacitor and LT1956. The easiest way to suppress input voltage transients is to add a small aluminum electrolytic capacitor in parallel with the low ESR input capacitor. The selected capacitor needs to have the right amount of ESR in order to critically dampen the resonant circuit formed by the input lead inductance and the input capacitor. The typical values of ESR will fall in the range of 0.5 to 2 and capacitance will fall in the range of 5F to 50F. If tantalum capacitors are used, values in the 22F to 470F range are generally needed to minimize ESR and meet ripple current and surge ratings. Care should be taken to ensure the ripple and surge ratings are not exceeded. The AVX TPS and Kemet T495 series are surge rated. AVX recommends derating capacitor operating voltage by 2 for high surge applications. CATCH DIODE Highest efficiency operation requires the use of a Schottky type diode. DC switching losses are minimized due to its low forward voltage drop, and AC behavior is benign due to its lack of a significant reverse recovery time. Schottky diodes are generally available with reverse voltage ratings of up to 60V and even 100V, and are price competitive with other types. The use of so-called "ultrafast" recovery diodes is generally not recommended. When operating in continuous mode, the reverse recovery time exhibited by "ultrafast" diodes will result in a slingshot type effect. The power
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internal switch will ramp up VIN current into the diode in an attempt to get it to recover. Then, when the diode has finally turned off, some tens of nanoseconds later, the VSW node voltage ramps up at an extremely high dV/dt, perhaps 5 to even 10V/ns ! With real world lead inductances, the VSW node can easily overshoot the VIN rail. This can result in poor RFI behavior and if the overshoot is severe enough, damage the IC itself. The suggested catch diode (D1) is an International Rectifier 10MQ060N Schottky. It is rated at 1.5A average forward current and 60V reverse voltage. Typical forward voltage is 0.63V at 1A. The diode conducts current only during switch off time. Peak reverse voltage is equal to regulator input voltage. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT (1 - DC) This formula will not yield values higher than 1.5A with maximum load current of 1.5A. The only reason to consider a larger diode is the worst-case condition of a high input voltage and shorted output. With a shorted condition, diode current will increase to a typical value of 2A, determined by peak switch current limit. This is safe for short periods of time, but it would be prudent to check with the diode manufacturer if continuous operation under these conditions must be tolerated. BOOST PIN For most applications, the boost components are a 0.1F capacitor and an MMSD914TI diode. The anode is typically connected to the regulated output voltage to generate a voltage approximately VOUT above VIN to drive the output stage. However, the output stage discharges the boost capacitor during the on time of the switch. The output driver requires at least 3V of headroom throughout this period to keep the switch fully saturated. If the output voltage is less than 3V, it is recommended that an alternate boost supply is used. The boost diode can be connected to the input, although, care must be taken to prevent the 2x VIN boost voltage from exceeding the BOOST pin absolute maximum rating. The additional voltage across the switch driver also increases power loss, reducing efficiency. If available, an independent supply can be used with a local bypass capacitor.
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A 0.1F boost capacitor is recommended for most applications. Almost any type of film or ceramic capacitor is suitable, but the ESR should be <1 to ensure it can be fully recharged during the off time of the switch. The capacitor value is derived from worst-case conditions of 1800ns on time, 42mA boost current and 0.7V discharge ripple. The boost capacitor value could be reduced under less demanding conditions, but this will not improve circuit operation or efficiency. Under low input voltage and low load conditions, a higher value capacitor will reduce discharge ripple and improve start-up operation. SHUTDOWN FUNCTION AND UNDERVOLTAGE LOCKOUT Figure 4 shows how to add undervoltage lockout (UVLO) to the LT1956. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur. Threshold voltage for lockout is about 2.38V. A 5.5A bias current flows out of the pin at this threshold. The internally generated current is used to force a default high state on the shutdown pin if the pin is left open. When low shutdown current is not an issue, the error due to this current can be minimized by making RLO 10k or less. If shutdown current is an issue, RLO can be raised to 100k, but the error due to initial bias current and changes with temperature should be considered.
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R LO = 10k to 100k (25k suggested) R HI = RLO ( VIN - 2.38 V ) 2.38 V - R LO(5.5 A)
VIN = minimum input voltage Keep the connections from the resistors to the shutdown pin short and make sure that interplane or surface capacitance to the switching nodes are minimized. If high resistor values are used, the shutdown pin should be
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LT1956 INPUT RHI SHDN IN
C2
RLO
bypassed with a 1000pF capacitor to prevent coupling problems from the switch node. If hysteresis is desired in the undervoltage lockout point, a resistor RFB can be added to the output node. Resistor values can be calculated from: R HI = RLO VIN - 2.38 ( V/ VOUT + 1) + V
[
R FB = (RHI ) VOUT /V
(
2.38 - RLO (5 .5A )
]
)
25k suggested for RLO VIN = Input voltage at which switching stops as input voltage descends to trip level V = Hysteresis in input voltage level Example: output voltage is 5V, switching is to stop if input voltage drops below 12V and should not restart unless input rises back to 13.5V. V is therefore 1.5V and VIN = 12V. Let RLO = 25k.
R HI =
2.24 R FB = 116k 5 / 1.5 = 387k
) 2.38 - 25k(5.5A ) 25k (10.41) = = 116k ( )
25k 12 - 2.38 1.5 / 5 + 1 + 1.5
[
(
]
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RFB L1 2.38V
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+
STANDBY
VSW
OUTPUT
5.5A
- +
C1
+
TOTAL SHUTDOWN 0.4V
-
GND
1956 F04
Figure 4. Undervoltage Lockout
SYNCHRONIZING The SYNC input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 10% and 90%. The input can be driven directly from a logic level output. The synchronizing range is equal to initial operating frequency up to 700kHz. This means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (570kHz), not the typical operating frequency of 500kHz. Caution should be used when synchronizing above 662kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs at input voltages less than twice output voltage. Higher inductor values will tend to eliminate this problem. See Frequency Compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensation. Application Note 19 has more details on the theory of slope compensation. At power-up, when VC is being clamped by the FB pin (see Figure 2, Q2), the sync function is disabled. This allows the frequency foldback to operate in the shorted output condition. During normal operation, switching frequency is controlled by the internal oscillator until the FB pin reaches 0.8V, after which the SYNC pin becomes operational. If no synchronization is required, this pin should be connected to ground.
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LAYOUT CONSIDERATIONS
As with all high frequency switchers, when considering layout, care must be taken in order to achieve optimal electrical, thermal and noise performance. For maximum efficiency, switch rise and fall times are typically in the nanosecond range. To prevent noise both radiated and conducted, the high speed switching current path, shown in Figure 5, must be kept as short as possible. This is implemented in the suggested layout of Figure 6. Shortening this path will also reduce the parasitic trace inductance of approximately 25nH/inch. At switch off, this parasitic inductance produces a flyback spike across the LT1956 switch. When operating at higher currents and input voltages, with poor layout, this spike can generate voltages across the LT1956 that may exceed its absolute maximum rating. A ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise.
L1 C1 MINIMIZE LT1956 C3-D1 LOOP GND D1 C2 GND SW SYNC LT1956 FB BOOST VC BIAS VIN GND GND RC CC KEEP FB AND VC COMPONENTS AWAY FROM HIGH FREQUENCY, HIGH CURRENT COMPONENTS
1956 F06
VIN C3
PLACE FEEDTHROUGH AROUND GROUND PINS (4 CORNERS) FOR GOOD THERMAL CONDUCTIVITY
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LT1956 L1 5V VIN C3 HIGH FREQUENCY CIRCULATING PATH D1 C1 LOAD
1956 F05
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Figure 5. High Speed Switching Path
The VC and FB components should be kept as far away as possible from the switch and boost nodes. The LT1956 pinout has been designed to aid in this. The ground for these components should be separated from the switch current path. Failure to do so will result in poor stability or subharmonic like oscillation.
CONNECT TO GROUND PLANE GND
D2 VOUT
FOR THE FE PACKAGE, SOLDER THE EXPOSED PAD TO THE COPPER GROUND PLANE UNDERNEATH THE DEVICE
GND
SHDN
KELVIN SENSE VOUT
R2 R1 CFB CF
Figure 6. Suggested Layout
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Board layout also has a significant effect on thermal resistance. For the GN package, Pins 1, 8, 9 and 16, GND, are a continuous copper plate that runs under the LT1956 die. This is the best thermal path for heat out of the package. Reducing the thermal resistance from Pins 1, 8, 9 and 16 onto the board will reduce die temperature and increase the power capability of the LT1956. This is achieved by providing as much copper area as possible around these pins. Adding multiple solder filled feedthroughs under and around these four corner pins to the ground plane will also help. Similar treatment to the catch diode and coil terminations will reduce any additional heating effects. For the FE package, the exposed pad should be soldered to the copper ground plane underneath the device. PARASITIC RESONANCE Resonance or "ringing" may sometimes be seen on the switch node (see Figure 7). Very high frequency ringing following switch rise time is caused by switch/diode/input capacitor lead inductance and diode capacitance. Schottky diodes have very high "Q" junction capacitance that can ring for many cycles when excited at high frequency. If total lead length for the input capacitor, diode and switch path is 1 inch, the inductance will be approximately 25nH. At switch off, this will produce a spike across the NPN output device in addition to the input voltage. At higher currents this spike can be in the order of 10V to 20V or higher with a poor layout, potentially exceeding the absolute max switch voltage. The path around switch, catch diode and input capacitor must be kept as short as possible to ensure reliable operation. When looking at this,
SW RISE
SW FALL
2V/DIV 0.2A/DIV INDUCTOR CURRENT AT IOUT = 0.1A
50ns/DIV
1956 F07
Figure 7. Switch Node Resonance
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a >100MHz oscilloscope must be used, and waveforms should be observed on the leads of the package. This switch off spike will also cause the SW node to go below ground. The LT1956 has special circuitry inside which mitigates this problem, but negative voltages over 0.8V lasting longer than 10ns should be avoided. Note that 100MHz oscilloscopes are barely fast enough to see the details of the falling edge overshoot in Figure 7. A second, much lower frequency ringing is seen during switch off time if load current is low enough to allow the inductor current to fall to zero during part of the switch off time (see Figure 8). Switch and diode capacitance resonate with the inductor to form damped ringing at 1MHz to 10 MHz. This ringing is not harmful to the regulator and it has not been shown to contribute significantly to EMI. Any attempt to damp it with a resistive snubber will degrade efficiency. THERMAL CALCULATIONS Power dissipation in the LT1956 chip comes from four sources: switch DC loss, switch AC loss, boost circuit current, and input quiescent current. The following formulas show how to calculate each of these losses. These formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. Switch loss:
PSW = RSW (IOUT ) ( VOUT ) + tEFF (1/ 2)(IOUT )( VIN)( f) VIN
2
10V/DIV SWITCH NODE VOLTAGE VIN = 25V VOUT = 5V L = 15H 500ns/DIV
1956 F08
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Figure 8. Discontinuous Mode Ringing
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Boost current loss:
VOUT2 (IOUT / 36) PBOOST = VIN
Quiescent current loss: PQ = VIN (0.0015) + VOUT (0.003) RSW = switch resistance ( 0.3) hot tEFF = effective switch current/voltage overlap time = (tr + tf + tIr + tIf) tr = (VIN/1.2)ns tf = (VIN/1.7)ns tIr = tIf = (IOUT/0.05)ns f = switch frequency Example: with VIN = 12V, VOUT = 5V and IOUT = 1A:
PSW
(0.3)(1)2 (5) + =
2
57 *10 -9 (1/ 2)(1)(12 ) 500 *10 3 12 = 0.125 + 0.171 = 0.296W
(
)
(
PBOOST =
12 PQ = 12(0.0015) + 5(0.003) = 0.033W
(5) (1 / 36) = 0.058W
Total power dissipation in the IC is given by: PTOT = PSW + PBOOST + PQ = 0.296W + 0.058W + 0.033W = 0.39W Thermal resistance for the LT1956 packages is influenced by the presence of internal or backside planes. SSOP (GN16) Package: With a full plane under the GN16 package, thermal resistance will be about 85C/W. TSSOP (Exposed Pad) Package: With a full plane under the TSSOP package, thermal resistance (JA) will be about 45C/W. To calculate die temperature, use the proper thermal resistance (JA) number for the desired package an add in worst-case ambient temperature: TJ = TA + (JA * PTOT) When estimating ambient, remember the nearby catch diode and inductor will also be dissipating power.
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PDIODE = ( VF )( VIN - VOUT )(ILOAD ) VIN
VF = Forward voltage of diode (assume 0.63V at 1A) PDIODE = (0.63 )(12 - 5 )(1) = 0.37W 12 Notice that the catch diode's forward voltage contributes a significant loss in the overall system efficiency. A larger, low VF diode can improve efficiency by several percent. PINDUCTOR = (ILOAD)(LDCR) LDCR = inductor DC resistance (assume 0.1) PINDUCTOR = (1)(0.1) = 0.1W Typical thermal resistance of the board is 10C/W. Taking the catch diode and inductor power dissipation into account and using the example calculations for LT1956 dissipation, the LT1956 die temperature will be estimated as: TJ = TA + (JA * PTOT) + (10 * [PDIODE + PINDUCTOR]) With the GN16 package (JA = 85C/W), at an ambient temperature of 70C: TJ = 70 + (85 * 0.39) + (10 * 0.47) = 108C With the TSSOP package (JA = 45C/W) at an ambient temperature of 70C: TJ = 70 + (45 * 0.37) + (10 * 0.47) = 91C Die temperature can peak for certain combinations of VIN, VOUT and load current. While higher VIN gives greater switch AC losses, quiescent and catch diode losses, a lower VIN may generate greater losses due to switch DC losses. In general, the maximum and minimum VIN levels should be checked with maximum typical load current for calculation of the LT1956 die temperature. If a more accurate die temperature is required, a measurement of the SYNC pin resistance (to GND) can be used. The SYNC pin resistance can be measured by forcing a voltage no greater than 0.5V at the pin and monitoring the pin current over temperature in a oven. This should be done with minimal device power (low VIN and no switching [VC = 0V]) in order to calibrate SYNC pin resistance with ambient (oven) temperature.
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Note: Some of the internal power dissipation in the IC, due to BOOST pin voltage, can be transferred outside of the IC to reduce junction temperature by increasing the voltage drop in the path of the boost diode D2 (see Figure 9). This reduction of junction temperature inside the IC will allow higher ambient temperature operation for a given set of conditions. BOOST pin circuitry dissipates power given by:
PDISS (BOOST Pin) = VOUT * (ISW / 36) * VC2 VIN
Typically, VC2 (the boost voltage across the capacitor C2) equals VOUT. This is because diodes D1 and D2 can be considered almost equal, where: VC2 = VOUT - VF(D2) - [-VF(D1)] = VOUT. Hence, the equation for boost circuitry power dissipation given in the previous Thermal Calculations section, is stated as:
PDISS(BOOST) VOUT * (ISW / 36) * VOUT = VIN
Here it can be seen that boost power dissipation increases as the square of VOUT. It is possible, however, to reduce VC2 below VOUT to save power dissipation by increasing the voltage drop in the path of D2. Care should be taken that VC2 does not fall below the minimum 3.3V boost voltage required for full saturation of the internal power switch. For output voltages of 5V, VC2 is approximately 5V. During switch turn on, VC2 will fall as the boost capacitor C2 is discharged by the BOOST pin. In the previous BOOST Pin section, the value of C2 was designed for a 0.7V droop in VC2 (= VDROOP). Hence, an output voltage as low as 4V would still allow the minimum 3.3V for the boost function using the C2 capacitor calculated. If a target output voltage of 12V is required, however, an excess of 8V is placed across the boost capacitor which is not required for the boost function but still dissipates additional power. What is required is a voltage drop in the path of D2 to achieve minimal power dissipation while still maintaining minimum boost voltage across C2.
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A zener, D4, placed in series with D2 (see Figure 9), drops voltage to C2. Example: The BOOST pin power dissipation for a 20V input to 12V output conversion at 1A is given by:
W
UU
PBOOST =
12 * (1 / 36) * 12 = 0.2W 20
If a 7V zener is placed in series with D2, then power dissipation becomes:
PBOOST =
12 * (1 / 36) * 5 = 0.084 W 20
For an FE package with thermal resistance of 45C/W, ambient temperature savings would be: T (ambient) savings = 0.116W * 45C/W = 5C For a GN package with thermal resistance of 85C/W, ambient temperature savings would be: T (ambient) savings = 0.116W * 85C/W = 10C The 7V zener should be sized for excess of 0.116W operation. The tolerances of the zener should be considered to ensure minimum VBOOST exceeds 3.3V + VDROOP.
D2 D4
D2
BOOST VIN C3 VIN SHDN SYNC GND LT1956 SW BIAS
C2
L1 VOUT
R1 FB VC D1 R2
+
C1
RC CC
CF
1956 F09
Figure 9. BOOST Pin, Diode Selection
1956f
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
Input Voltage vs Operating Frequency Considerations The absolute maximum input supply voltage for the LT1956 is specified at 60V. This is based on internal semiconductor junction breakdown effects. The practical maximum input supply voltage for the LT1956 may be less than 60V due to internal power dissipation or switch minimum on time considerations. For the extreme case of an output short-circuit fault to ground, see the section Short-Circuit Considerations. A detailed theoretical basis for estimating internal power dissipation is given in the Thermal Calculations section. This will allow a first pass check of whether an application's maximum input voltage requirement is suitable for the LT1956. Be aware that these calculations are for DC input voltages and that input voltage transients as high as 60V are possible if the resulting increase in internal power dissipation is of insufficient time duration to raise die temperature significantly. For the FE package, this means high voltage transients on the order of hundreds of milliseconds are possible. If LT1956 (FE package) thermal calculations show power dissipation is not suitable for the given application, the LT1766 (FE package) is a recommended alternative since it is identical to the LT1956 but runs cooler at 200kHz. Switch minimum on time is the other factor that may limit the maximum operational input voltage for the LT1956 if pulse-skipping behavior is not allowed. For the LT1956, pulse-skipping may occur for VIN/(VOUT + VF) ratios > 4. (VF = Schottky diode D1 forward voltage drop, Figure 5.) If the LT1766 is used, the ratio increases to 10. Pulseskipping is the regulator's way of missing switch pulses to maintain output voltage regulation. Although an increase in output ripple voltage can occur during pulse-skipping, a ceramic output capacitor can be used to keep ripple voltage to a minimum (see output ripple voltage comparison for tantalum vs ceramic output capacitors, Figure 3). FREQUENCY COMPENSATION Before starting on the theoretical analysis of frequency response, the following should be remembered--the worse the board layout, the more difficult the circuit will be to stabilize. This is true of almost all high frequency analog
U
circuits, read the Layout Considerations section first. Common layout errors that appear as stability problems are distant placement of input decoupling capacitor and/ or catch diode, and connecting the VC compensation to a ground track carrying significant switch current. In addition, the theoretical analysis considers only first order non-ideal component behavior. For these reasons, it is important that a final stability check is made with production layout and components. The LT1956 uses current mode control. This alleviates many of the phase shift problems associated with the inductor. The basic regulator loop is shown in Figure 10. The LT1956 can be considered as two gm blocks, the error amplifier and the power stage. Figure 11 shows the overall loop response. At the VC pin, the frequency compensation components used are: RC = 2.2k, CC = 0.022F and CF = 220pF. The output capacitor used is a 100F, 10V tantalum capacitor with typical ESR of 100m. The ESR of the tantalum output capacitor provides a useful zero in the loop frequency response for maintaining stability. This ESR, however, contributes significantly to the ripple voltage at the output (see Output Ripple Voltage in the Applications Information section). It is possible to reduce capacitor size and output ripple voltage by replacing the tantalum output capacitor with a ceramic output capacitor because of its very low ESR. The zero provided by the tantalum output capacitor must now be reinserted back into the loop. Alternatively, there may be cases where, even with the tantalum output capacitor, an additional zero is required in the loop to increase phase margin for improved transient response. A zero can be added into the loop by placing a resistor (RC) at the VC pin in series with the compensation capacitor, CC, or by placing a capacitor (CFB) between the output and the FB pin. When using RC, the maximum value has two limitations. First, the combination of output capacitor ESR and RC may stop the loop rolling off altogether. Second, if the loop gain is not rolled off sufficiently at the switching frequency, output ripple will perturb the VC pin enough to cause unstable duty cycle switching similar to subharmonic
1956f
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21
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
LT1956 CURRENT MODE POWER STAGE gm = 2mho SW ERROR AMPLIFIER FB gm = 2000mho RO 200k GND VC R2 RC CC
TANTALUM ESR RLOAD
CERAMIC ESL C1
GAIN (dB)
CFB
R1
1.22V
CF
1956 F10
Figure 10. Model for Loop Response
oscillations. If needed, an additional capacitor (CF) can be added across the RC/CC network from the VC pin to ground to further suppress VC ripple voltage. With a tantalum output capacitor, the LT1956 already includes a resistor (RC) and filter capacitor (CF) at the VC pin (see Figures 10 and 11) to compensate the loop over the entire VIN range (to allow for stable pulse skipping for high VIN-to-VOUT ratios 4). A ceramic output capacitor can still be used with a simple adjustment to the resistor RC for stable operation (see Ceramic Capacitors section for stabilizing LT1956). If additional phase margin is required, a capacitor (CFB) can be inserted between the output and FB pin but care must be taken for high output voltage applications. Sudden shorts to the output can create unacceptably large negative transients on the FB pin. For VIN-to-VOUT ratios < 4, higher loop bandwidths are possible by readjusting the frequency compensation components at the VC pin. When checking loop stability, the circuit should be operated over the application's full voltage, current and temperature range. Proper loop compensation may be obtained by empirical methods as described in Application Notes 19 and 76.
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80 60 180 150 GAIN 40 20 PHASE 0 -20 -40 60 30 0 1M
1956 F11
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- +
OUTPUT
120 90
PHASE (DEG)
+
C1
10
1k 10k 100k FREQUENCY (Hz) VIN = 12V RC = 2.2k VOUT = 5V CC = 22nF ILOAD = 500mA CF = 220pF COUT = 100F, 10V, 0.1
100
Figure 11. Overall Loop Response
CONVERTER WITH BACKUP OUTPUT REGULATOR In systems with a primary and backup supply, for example, a battery powered device with a wall adapter input, the output of the LT1956 can be held up by the backup supply with the LT1956 input disconnected. In this condition, the SW pin will source current into the VIN pin. If the SHDN pin is held at ground, only the shut down current of 25A will be pulled via the SW pin from the second supply. With the SHDN pin floating, the LT1956 will consume its quiescent operating current of 1.5mA. The VIN pin will also source current to any other components connected to the input line. If this load is greater than 10mA or the input could be shorted to ground, a series Schottky diode must be added, as shown in Figure 12. With these safeguards, the output can be held at voltages up to the VIN absolute maximum rating. BUCK CONVERTER WITH ADJUSTABLE SOFT-START Large capacitive loads or high input voltages can cause high input currents at start-up. Figure 13 shows a circuit that limits the dv/dt of the output at start-up, controlling the capacitor charge rate. The buck converter is a typical configuration with the addition of R3, R4, CSS and Q1. As the output starts to rise, Q1 turns on, regulating switch
1956f
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
D3 10MQ060N REMOVABLE INPUT
VIN R3 54k SHDN SYNC GND R4 25k C3 2.2F
Figure 12. Dual Source Supply with 25A Reverse Leakage
BOOST INPUT 12V C3 2.2F CERAMIC VIN
LT1956 SHDN SYNC GND FB VC Q1 RC 2.2k CC 0.022F CF 220pF
Figure 13. Buck Converter with Adjustable Soft-Start
current via the VC pin to maintain a constant dv/dt at the output. Output rise time is controlled by the current through CSS defined by R4 and Q1's VBE. Once the output is in regulation, Q1 turns off and the circuit operates normally. R3 is transient protection for the base of Q1.
RiseTime =
(R4)(CSS )(VOUT )
VBE
Using the values shown in Figure 10, Rise Time =
(47 * 103 )(15 * 10-9 )(5) = 5ms
0.7
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MMSD914TI C2 0.1F SW BIAS R1 15.4k FB VC RC 2.2k CC 0.022F CF 220pF
1956 F12
W
UU
BOOST LT1956
L1 18H 5V, 1A ALTERNATE SUPPLY
D1 10MQ060N
R2 4.99k
+
C1 100F 10V
D2 MMSD914TI C2 0.1F
BIAS SW
L1 18H D1 C1 100F
+
R1 15.4k
OUTPUT 5V 1A
R2 4.99k R3 2k R4 47k CSS 15nF
1766 F13
The ramp is linear and rise times in the order of 100ms are possible. Since the circuit is voltage controlled, the ramp rate is unaffected by load characteristics and maximum output current is unchanged. Variants of this circuit can be used for sequencing multiple regulator outputs. DUAL POLARITY OUTPUT CONVERTER The circuit in Figure 14a generates both positive and negative 5V outputs with all components under 3mm height. The topology for the 5V output is a standard buck converter. The -5V output uses a second inductor L2, diode D3 and output capacitor C6. The capacitor C4
1956f
23
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
VIN 9V TO 12V (TRANSIENTS TO 36V) C3 2.2F 50V CERAMIC
VIN LT1956 SHDN SYNC GND RC 2.2k CC 3300pF
GND C4 10F *SUMIDA CDRH4D28-150 6.3V **SEE FIGURE 14c FOR VOUT1, VOUT2 CER LOAD CURRENT RELATIONSHIP IF LOAD CAN GO TO ZERO, AN OPTIONAL PRELOAD OF 500 CAN BE USED TO IMPROVE REGULATION
Figure 14a. Dual Polarity Output Converter
500
VOUT2 MAXIMUM LOAD CURRENT (mA)
5.30 5.25 5.20 5.15 |VOUT2| (V) 5.10 5.05 5.00 4.95 4.90 4.85 4.80 4.75 0 600 400 200 VOUT1 LOAD CURRENT (mA) 800
1956 F15b
450 400 350 300 250 200 150 100 50 0
EFFICIENCY (%)
0
Figure 14b. VOUT2 (-5V) Maximum Allowable Load Current vs VOUT1 (5V) Load Current
couples energy to L2 and ensures equal voltages across L2 and L1 during steady state. Instead of using a transformer for L1 and L2, uncoupled inductors were used because they require less height than a single transformer, can be placed separately in the circuit layout for optimized space savings and reduce overall cost. This is true even when the uncoupled inductors are sized (twice the value of inductance of the transformer) in order to keep ripple current comparable to the transformer solution. If a single
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U
D2 MMSD914TI C2 0.1F BOOST SW R1 15.4k FB VC CF 220pF R2 4.99k D1 B0540W L1* 15H VOUT1** 5V
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UU
+
C5 10F 6.3V CER
+
L2*
C6 10F 6.3V CER
+
VOUT2** -5V
D3 B0540W
1956 F14a
100
VOUT1 LOAD CURRENT 750mA VOUT1 LOAD CURRENT 500mA VOUT1 LOAD CURRENT 250mA
90 80 70 60 50 40 30 20 10
VOUT1 LOAD CURRENT 750mA VOUT1 LOAD CURRENT 250mA
100 200 300 400 500 VOUT2 LOAD CURRENT (mA)
600
0
0
100 300 400 200 VOUT2 LOAD CURRENT (mA)
500
1956 F14d
1956 F14c
Figure 14c. VOUT2 (-5V) Output Voltage vs Load Current
Figure 14d. Dual Polarity Output Converter Efficiency
transformer becomes available to provide a better height/ cost solution, refer to the dual output SEPIC circuit description in Design Note 100 for correct transformer connection. During switch on-time, in steady state, the voltage across both L1 and L2 is positive and equal; with energy (and current) ramping up in each inductor. The current in L2 is provided by the coupling capacitor C4. During switch offtime, current ramps downward in each inductor. The
1956f
LT1956/LT1956-5
APPLICATIO S I FOR ATIO
current in L2 and C4 flows via the catch diode D3, charging the negative output capacitor C6. If the negative output is not loaded enough, it can go severely unregulated (become more negative). Figure 14b shows the maximum allowable -5V output load current (vs load current on the 5V output) that will maintain the -5V output within 3% tolerance. Figure 14c shows the -5V output voltage regulation vs its own load current when plotted for three separate load currents on the 5V output. The efficiency of the dual output converter circuit shown in Figure 14a is given in Figure 14d. POSITIVE-TO-NEGATIVE CONVERTER The circuit in Figure 15 is a positive-to-negative topology using a grounded inductor. It differs from the standard approach in the way the IC chip derives its feedback signal because the LT1956 accepts only positive feedback signals. The ground pin must be tied to the regulated negative output. A resistor divider to the FB pin, then provides the proper feedback voltage for the chip. The following equation can be used to calculate maximum load current for the positive-to-negative converter:
D2 MMSD914TI C2 0.1F VIN 12V C3 2.2F 25V BOOST VIN LT1956 GND VC FB D1 10MQO60N SW R1 36.5k
L1* 7H
+
R2 4.12k
CC CF RC
* INCREASE L1 TO 10H OR 18H FOR HIGHER CURRENT APPLICATIONS. SEE APPLICATIONS INFORMATION ** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION
Figure 15. Positive-to-Negative Converter
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IMAX (VIN )(VOUT ) IP - 2(V (VOUT )(VIN - 0.3) OUT + VIN )(f)(L) = (VOUT + VIN - 0.3)(VOUT + VF )
IP = maximum rated switch current VIN = minimum input voltage VOUT = output voltage VF = catch diode forward voltage 0.3 = switch voltage drop at 1.5A Example: with VIN(MIN) = 5.5V, VOUT = 12V, L = 15H, VF = 0.63V, IP = 1.5A: IMAX = 0.36A. INDUCTOR VALUE The criteria for choosing the inductor is typically based on ensuring that peak switch current rating is not exceeded. This gives the lowest value of inductance that can be used, but in some cases (lower output load currents) it may give a value that creates unnecessarily high output ripple voltage. The difficulty in calculating the minimum inductor size needed is that you must first decide whether the switcher will be in continuous or discontinuous mode at the critical point where switch current reaches 1.5A. The first step is to use the following formula to calculate the load current above which the switcher must use continuous mode. If your load current is less than this, use the discontinuous mode formula to calculate minimum inductor needed. If load current is higher, use the continuous mode formula. Output current where continuous mode is needed:
ICONT > ( VIN )2 (IP )2 4( VIN + VOUT )( VIN + VOUT + VF )
C1 100F 20V TANT OUTPUT** -12V, 0.25A
1956 F15
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Minimum inductor discontinuous mode: LMIN = 2( VOUT )(IOUT ) ( f)(IP )2
1956f
25
LT1956/LT1956-5
PACKAGE DESCRIPTIO
LMIN =
Minimum inductor continuous mode:
(VIN )(VOUT ) (V + VF ) 2(f)(VIN + VOUT )IP - IOUT 1 + OUT VIN
For a 12V to -12V converter using the LT1956 with peak switch current of 1.5A and a catch diode of 0.63V:
ICONT > (12)2 (1.5)2 = 0.370 A 4(12 + 12)(12 + 12 + 0.63)
For a load current of 0.25A, this says that discontinuous mode can be used and the minimum inductor needed is found from: 2(12)(0.25) LMIN = = 5.3H (500 * 103 )(1.5)2 In practice, the inductor should be increased by about 30% over the calculated minimum to handle losses and variations in value. This suggests a minimum inductor of 7H for this application. Ripple Current in the Input and Output Capacitors Positive-to-negative converters have high ripple current in the input capacitor. For long capacitor lifetime, the RMS value of this current must be less than the high frequency ripple current rating of the capacitor. The following formula will give an approximate value for RMS ripple current. This formula assumes continuous mode and large inductor value. Small inductors will give somewhat higher ripple current, especially in discontinuous mode. The exact formulas are very complex and appear in Application Note 44, pages 29 and 30. For our purposes here I have simply added a fudge factor (ff). The value for ff is about 1.2 for higher load currents and L 15H. It increases to about 2.0 for smaller inductors at lower load currents.
Capacitor IRMS = ( ff)(IOUT ) VOUT VIN
ff = 1.2 to 2.0
26
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The output capacitor ripple current for the positive-tonegative converter is similar to that for a typical buck regulator--it is a triangular waveform with peak-to-peak value equal to the peak-to-peak triangular waveform of the inductor. The low output ripple design in Figure 14 places the input capacitor between VIN and the regulated negative output. This placement of the input capacitor significantly reduces the size required for the output capacitor (versus placing the input capacitor between VIN and ground). The peak-to-peak ripple current in both the inductor and output capacitor (assuming continuous mode) is: IP-P = DC * VIN f *L VOUT + VF VOUT + VIN + VF DC = Duty Cycle = ICOUT (RMS) = IP-P 12 The output ripple voltage for this configuration is as low as the typical buck regulator based predominantly on the inductor's triangular peak-to-peak ripple current and the ESR of the chosen capacitor (see Output Ripple Voltage in Applications Information). Diode Current
Average diode current is equal to load current. Peak diode current will be considerably higher.
Peak diode current:
Continuous Mode = (V + V ) ( VIN )( VOUT ) IOUT IN OUT + VIN 2(L)( f)( VIN + VOUT ) Discontinuous Mode = 2(IOUT )( VOUT ) (L)( f)
Keep in mind that during start-up and output overloads, average diode current may be much higher than with normal loads. Care should be used if diodes rated less than 1A are used, especially if continuous overload conditions must be tolerated.
1956f
LT1956/LT1956-5
PACKAGE DESCRIPTIO U
FE Package 16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BB
4.90 - 5.10* (.193 - .201) 3.58 (.141) 16 1514 13 12 1110 6.60 0.10 4.50 0.10 SEE NOTE 4 0.45 0.05 1.05 0.10 0.65 BSC RECOMMENDED SOLDER PAD LAYOUT 4.30 - 4.50* (.169 - .177) 0 - 8 12345678 1.10 (.0433) MAX 9
3.58 (.141)
2.94 (.116) 2.94 6.40 (.116) BSC
0.09 - 0.20 (.0036 - .0079)
0.45 - 0.75 (.018 - .030)
0.65 (.0256) BSC
NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE
0.195 - 0.30 (.0077 - .0118)
0.05 - 0.15 (.002 - .006)
FE16 (BB) TSSOP 0203
4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE
1956f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LT1956/LT1956-5
PACKAGE DESCRIPTIO U
GN Package 16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.045 .005 .189 - .196* (4.801 - 4.978) 16 15 14 13 12 11 10 9 .009 (0.229) REF .150 - .165 .229 - .244 (5.817 - 6.198) .0165 .0015 .150 - .157** (3.810 - 3.988) .0250 TYP 1 .015 .004 x 45 (0.38 0.10) .007 - .0098 (0.178 - 0.249) .016 - .050 (0.406 - 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0 - 8 TYP .053 - .068 (1.351 - 1.727) 23 4 56 7 8 .004 - .0098 (0.102 - 0.249) .008 - .012 (0.203 - 0.305) .0250 (0.635) BSC
GN16 (SSOP) 0502
.254 MIN
RECOMMENDED SOLDER PAD LAYOUT
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS Up to 64V Input, 100kHz, 5A and 2A Up to 75V Input, 60kHz Operation Up to 42V, 6A, 500kHz Switch Up to 35V, 3A, 500kHz Switch Operation Up to 25V Input, Synchronizable (LT1375), N8, S8, S16 3.6V to 25V VIN, 6-Lead ThinSOTTM 7.4V to 60V VIN, 100kHz Operation, 700mA Internal Switch, S8 VIN: 3V to 25V; VREF = 1.2V; S8, TSSOP-16E Exposed Pad 5.5V to 60V Input, 200kHz Operation, 1.5A Internal Switch, TSSOP-16E VIN: 3V to 25V; VREF = 1.2V; MS8 Up to 7.4V to 60V, 200kHz Operation, 700mA Internal Switch, TSSOP-16E Operation Up to 48V, Controlled Voltage and Current Slew Rates, S16
1956f
LT1074/LT1076/ Step-Down Switching Regulators LT1076HV LT1082 LT1370 LT1371 LT1375/LT1376 LT1616 LT1676 LT1765 LT1766 LT1767 LT1776 LT1777 1A High Voltage/Efficiency Switching Voltage Regulator High Efficiency DC/DC Converter High Efficiency DC/DC Converter 1.5A, 500kHz Step-Down Switching Regulators 600mA, 1.4MHz Step-Down Switching Regulator Wide Input Range, High Efficiency, Step-Down Switching Regulator Monolithic 3A, 1.25MHz Step-Down Regulator Wide Input Range, High Efficiency, Step-Down Switching Regulator Monolithic 1.5A, 1.25MHz Step-Down Regulator Wide Input Range, High Efficiency, Step-Down Switching Regulator Low Noise Buck Regulator
ThinSOT is a trademark of Linear Technology Corporation.
28
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507
q
LT/TP 0303 2K * PRINTED IN USA
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